1766 查看數據表（PDF） - Linear Technology

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1766

5.5V to 60V 1.5A, 200kHz Step-Down Switching Regulator

Linear Technology 

LT1766/LT1766-5

APPLICATIONS INFORMATION

Discontinuous mode is entered when the output load

current is less than one-half of the inductor ripple current

(ILP-P). In this mode, inductor current falls to zero before

the next switch turn on (see Figure 8). Buck converters

will be in discontinuous mode for output load current

given by:

IOUT

< (VOUT + VF )(VIN – VOUT – VF )

Discontinuous Mode

(2)(VIN)(f)(L)

The inductor value in a buck converter is usually chosen

large enough to keep inductor ripple current (ILP-P) low;

this is done to minimize output ripple voltage and maximize

output load current. In the case of large inductor values,

as seen in the equation above, discontinuous mode will

be associated with light loads.

When choosing small inductor values, however, discon-

tinuous mode will occur at much higher output load cur-

rents. The limit to the smallest inductor value that can be

chosen is set by the LT1766 peak switch current (IP) and

the maximum output load current required, given by:

IOUT(MAX)

Discontinuous

Mode

=

IP2

(2)(ILP-P )

=

(IP )2 ((f)(L)(VIN))

2(VOUT + VF )(VIN – VOUT – VF )

Example: For VIN = 15V, VOUT = 5V, VF = 0.63V, f = 200kHz

and L = 10μH.

IOUT(MAX)

Discontinuous

Mode

= (1.5)2 • (200 •103)(10–5)(15)

2(5 + 0.63)(15 – 5 – 0.63)

IOUT(MAX)

= 0.639A

Discontinuous Mode

What has been shown here is that if high inductor ripple

current and discontinuous mode operation can be tolerated,

small inductor values can be used. If a higher output load

current is required, the inductor value must be increased.

If IOUT(MAX) no longer meets the discontinuous mode

criteria, use the IOUT(MAX) equation for continuous mode;

the LT1766 is designed to operate well in both modes of

operation, allowing a large range of inductor values to

be used.

14

Short-Circuit Considerations

The LT1766 is a current mode controller. It uses the VC

node voltage as an input to a current comparator which

turns off the output switch on a cycle-by-cycle basis as

this peak current is reached. The internal clamp on the VC

node, nominally 2V, then acts as an output switch peak

current limit. This action becomes the switch current limit

specification. The maximum available output power is then

determined by the switch current limit.

A potential controllability problem could occur under

short-circuit conditions. If the power supply output is

short circuited, the feedback amplifier responds to the

low output voltage by raising the control voltage, VC,

to its peak current limit value. Ideally, the output switch

would be turned on, and then turned off as its current

exceeded the value indicated by VC. However, there is finite

response time involved in both the current comparator and

turn-off of the output switch. These result in a minimum

on-time, tON(MIN). When combined with the large ratio of

VIN to (VF + I • R), the diode forward voltage plus inductor

I • R voltage drop, the potential exists for a loss of control.

Expressed mathematically the requirement to maintain

control is:

f

•

tON

≤

VF

+I•R

VIN

where:

f = Switching frequency

tON = Switch minimum on-time

VF = Diode forward voltage

VIN = Input voltage

I • R = Inductor I • R voltage drop

If this condition is not observed, the current will not be

limited at IPK, but will cycle-by-cycle ratchet up to some

higher value. Using the nominal LT1766 clock frequency

of 200KHz, a VIN of 40V and a (VF + I • R) of say 0.7V, the

maximum tON to maintain control would be approximately

90ns, an unacceptably short time.

The solution to this dilemma is to slow down the oscil-

lator when the FB pin voltage is abnormally low thereby

indicating some sort of short-circuit condition. Oscillator

frequency is unaffected until FB voltage drops to about

2/3 of its normal value. Below this point the oscillator

1766fc

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