SC4502 查看數據表（PDF） - Semtech Corporation
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SC4502 Datasheet PDF : 19 Pages
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SC4502/SC4502H
POWER MANAGEMENT
Applications Information
The absolute maximum operating frequency of the
converter is therefore DMIN = 0.25 = 1.67MHz . The
150ns 150ns
actual operating frequency needs to be lower to allow
for modulating headroom.
The power transistor inside the SC4502/SC4502H is
turned off every switching cycle for an interval determined
by the discharge time of the oscillator ramp plus the
propagation delay of the power switch. This minimum off
time limits the maximum duty cycle of the regulator at a
given switching frequency. A boost converter with high
VOUT
VIn ratio requires long switch on time and high duty cycle.
If the required duty cycle is higher than the attainable
maximum, the converter will operate in dropout. (Dropout
is the condition in which the regulator cannot attain its
set output voltage below current limit.)
The minimum off times of closed-loop boost converters set
to various output voltages were measured by lowering their
input voltages until dropout occurs. It was found that the
minimum off time of the SC4502/SC4502H ranged from
80ns to 110ns at room temperature.
Beware of dropout while operating at very low input
voltages (1.5V-2V) with off time approaching 110ns.
Shorten the PCB trace between the power source and
the device input pin, as line drop may be a significant
percentage of the input voltage. A regulator in dropout
may appear as if it is in current limit. The cycle-by-cycle
current limit of the SC4502/SC4502H is duty-cycle and
input voltage invariant and is typically 2A. If the switch
current limit is not at least 1.4A, then the converter is
likely in dropout. The switching frequency should then be
lowered to improve controllability.
operating in continuous-conduction mode is
∆IL
=
( ) D⋅ VIN − VCESAT
f ⋅L
(5)
where f is the switching frequency and L is the inductance.
Substituting (3) into (5) and neglecting V ,
CESAT
∆IL
=
VIN
f ⋅L
1−
VIN
VOUT +
VD
(6)
In peak current-mode control, the slope of the modulating
(sensed switch current) ramp should be steep enough to
lessen jittery tendency but not so steep that large flux
swing decreases efficiency. Inductor ripple current DI
L
between 25%-40% of the peak inductor current limit is a
good compromise. Inductors so chosen are optimized in
size and DCR. Setting ∆IL = 0.3•(1.4A) = 0.42A, VD=0.5V
in (6),
L=
VIN
f ⋅ ∆IL
1 −
VIN
VOUT +
VD
=
VIN
0.42A
⋅
f
1
−
VOUT
VIN
+ 0.5V
(7)
where L is in µH and f is in MHz.
Equation (6) shows that for a given VOUT, ∆IL is the highest
when
VIN
=
(VOUT +
2
VD ) . If VIN varies over a
wide range,
then
choose L based on the nominal input voltage.
The saturation current of the inductor should be 20%-
30% higher than the peak current limit (2A). Low-cost
powder iron cores are not suitable for high-frequency
switching power supplies due to their high core losses.
Inductors with ferrite cores should be used.
Input Capacitor
Both the minimum on time and the minimum off time
reduce control range of the PWM regulator. Bench
measurement showed that reduced modulating range
started to be a problem at frequencies over 2MHz. Although
the oscillator is capable of running well above 2MHz,
controllability limits the maximum operating frequency.
The input current in a boost converter is the inductor
current, which is continuous with low RMS current ripples.
A 2.2µF-4.7µF ceramic input capacitor is adequate for
most applications.
Output Capacitor
Inductor Selection
The inductor ripple current ∆IL of a boost converter
Both ceramic and low ESR tantalum capacitors can be
used as output filtering capacitors. Multi-layer ceramic
capacitors, due to their extremely low ESR (<5mΩ), are
the best choice. Use ceramic capacitors with stable
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