MAX17006BETP 查看數據表（PDF） - Maxim Integrated

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MAX17006BETP

1.2MHz, Low-Cost, High-Performance Chargers

Maxim Integrated 

1.2MHz, Low-Cost,

High-Performance Chargers

The following additional loss occurs in the low-side

MOSFET due to the body diode conduction losses:

PDBDY (LS) = 0.05 × IPEAK × 0.4V

The total power low-side MOSFET dissipation is:

PDTOTAL (LS) ≈ PDCOND(LS) + PDBDY (LS)

These calculations provide an estimate and are not a

substitute for breadboard evaluation, preferably including

a verification using a thermocouple mounted on the

MOSFET.

Inductor Selection

The selection of the inductor has multiple trade-offs

between efficiency, transient response, size, and cost.

Small inductance is cheap and small, and has a better

transient response due to higher slew rate; however, the

efficiency is lower because of higher RMS current. High

inductance results in lower ripple so that the need of the

output capacitors for output-voltage ripple goes low.

The MAX17005B/MAX17006B/MAX17015B combine all

the inductor trade-offs in an optimum way by controlling

switching frequency. High-frequency operation permits

the use of a smaller and cheaper inductor, and conse-

quently results in smaller output ripple and better tran-

sient response.

The charge current, ripple, and operating frequency

(off-time) determine the inductor characteristics. For

optimum efficiency, choose the inductance according

to the following equation:

L=

k × VIN2

4 × ICHG × LIRMAX

where k = 35ns/V.

For optimum size and inductor current ripple, choose

LIRMAX = 0.4, which sets the ripple current to 40% the

charge current and results in a good balance between

inductor size and efficiency. Higher inductor values

decrease the ripple current. Smaller inductor values

save cost but require higher saturation current capabili-

ties and degrade efficiency.

Inductor L1 must have a saturation current rating of at

least the maximum charge current plus 1/2 the ripple

current (ΔIL):

ISAT = ICHG + (1/2) ΔIL

The ripple current is determined by:

ΔIL

=

k

× VIN2

4L

Input Capacitor Selection

The input capacitor must meet the ripple current

requirement (IRMS) imposed by the switching currents.

Nontantalum chemistries (ceramic, aluminum, or

OS-CON) are preferred due to their resilience to power-

up and surge currents:

( ) ⎛

IRMS = ICHG × ⎜

VBATT ×

VDCIN - VBATT

⎞

⎟

⎝⎜

VDCIN

⎠⎟

The input capacitors should be sized so that the tem-

perature rise due to ripple current in continuous conduc-

tion does not exceed approximately 10°C. The

maximum ripple current occurs at 50% duty factor or

VDCIN = 2 x VBATT, which equates to 0.5 x ICHG. If the

application of interest does not achieve the maximum

value, size the input capacitors according to the worst-

case conditions.

Output Capacitor Selection

The output capacitor absorbs the inductor ripple cur-

rent and must tolerate the surge current delivered from

the battery when it is initially plugged into the charger.

As such, both capacitance and ESR are important

parameters in specifying the output capacitor as a filter

and to ensure the stability of the DC-DC converter (see

the Compensation section.) Beyond the stability

requirements, it is often sufficient to make sure that the

output capacitor’s ESR is much lower than the battery’s

ESR. Either tantalum or ceramic capacitors can be

used on the output. Ceramic devices are preferable

because of their good voltage ratings and resilience to

surge currents. Choose the output capacitor based on:

COUT

=

fSW

IRIPPLE

× 8 × ΔVBATT

× kCAP−BIAS

Choose kCAP-BIAS is a derating factor of 2 for typical 25V-

rated ceramic capacitors.

For fSW = 800kHz, IRIPPLE = 1A, and to get ΔVBATT =

70mV, choose COUT as 4.7μF.

If the internal resistance of battery is close to the ESR of

the output capacitor, the voltage ripple is shared with

the battery and is less than calculated.

18 ______________________________________________________________________________________

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